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The latest generation of Virtex-II Pro™
and Virtex-II Pro X™ devices features
RocketIO™ and RocketIO X transceivers
that can drive high-speed serial links at
line rates of up to 10 Gbps. Two important
features of high-speed serial links
make the behavior of these signals very
different from those found on traditional
on-board buses. First are the shorter rise
time and associated higher bandwidth signals;
this makes the signals more sensitive
to small imperfections. Second are the
longer interconnect lengths; this makes
the signals more sensitive to attenuation
effects. Both effects contribute to rise time
degradation, inter-symbol interference
(ISI), and collapse of the eye diagram.
Although it is possible (and important)
to model and simulate these two physical
features, it is difficult to do so accurately.
We are still low on the learning curve,
where feedback from measurements on
real systems is critically important to
improve models and optimize the design
for performance.
When first article hardware is available,
measurements on the passive interconnects
can provide valuable insight on
the expected system-level performance
independent of your choice of silicon
drivers and receivers. With accurate
measurement-based models, you can
optimize the cost/performance tradeoffs
of silicon selection.
The Bandwidth of the Measurement
Bandwidth is the highest sine wave frequency
component that is significant.
“Significant” means the frequency at
which a harmonic of the signal is greater
than -3 dB of the amplitude the same harmonic
an ideal square wave at the same
clock frequency would have.
If the signal edge is roughly Gaussian
with a 10-90% rise time (RT), the bandwidth
(BW) is approximately:
BW = 0.35/RT
For example, a rise time of 0.1 ns has a
bandwidth of about 0.35/0.1 ~ 3.5 GHz.
Usually, the bit rate is specified in a highspeed
serial link. To estimate the bandwidth
of the signal, we need to have an estimate of
the rise time. Assuming that the rise time is
25% of the bit period, then the bandwidth
of the signal is approximately:
BWsignal = 0.35/0.25 BR~1.4 x BR
As a general rule of thumb, the highest
sine wave frequency component in a highspeed
serial link is about 1.4 times the bit
rate. For a 2.5 Gbps signal, the bandwidth
is about 3.5 GHz. If it is important to
know whether the bandwidth is really 3.5
GHz or 4 GHz, the term “bandwidth” is
misused, as it is not accurate enough to
make this fine a distinction. Rather, you
should use the entire spectrum.
To have confidence in the accuracy of a
model, the bandwidth of that model – the
highest sine wave frequency at which the
simulated electrical performance still
matches the measured performance of the
real structure – should be at least twice the
bandwidth of the signal to allow for a reasonable
margin. Likewise, the bandwidth
of the measurement should be at least twice
the bandwidth of the signal. This rule of
thumb suggests that the bandwidth of the
measurement should be at least:
BWmeasurement = 3 x BR
If the bit rate is 10 Gbps, the bandwidth
of any model used (or the bandwidth of the
measurement of the interconnect) should be
at least 30 GHz. Of course, if the rise time of
the bit pattern is longer than 25% of the
bit period, the measurement bandwidth
might be reduced from this rule of thumb.
Unfortunately, the higher the bandwidth
required, the more expensive it is
(both in resources, time, and money) to
perform a measurement or create a model
of an interconnect. That is why it is so
important to have a rough idea of the
bandwidth requirements so as to minimize
the cost. As high-speed serial links
approach the 10 Gbps rate, measurement
bandwidths need to be at least 30 GHz.
Accurate measurements in this regime get
increasingly more difficult with each generation
of bit rate.
No Such Thing as a Free Launch
Credit that clever turn of phrase to Scott
McMorrow, president of Teraspeed
Consulting. Probing a channel on a board
or a backplane introduces errors that might
not be there, or be of a different magnitude,
than in the actual product when signals
are launched from chips in packages.
All high-performance measurement
instruments, such as a time domain reflectometer
(TDR) or a vector network analyzer
(VNA), have a standard connector on
the front face, typically APC-7 or 3.5 mm.
High-performance cables are used to get
from the instrument to the device under
test. However, the interface from the cable
to the board traces under test can introduce
impedance discontinuities which degrade
the signal getting onto the trace.
The larger the discontinuity, the more
high-frequency components reflect back
to the source, and the fewer that get
launched into the transmission line. If
characterizing a path for 5 Gbps signals,
the connection method may limit the
measured system performance. To increase
the bandwidth of the characterization,
you must consider the launch before
designing and building the board.
A key ingredient in the design for test
for high-bandwidth characterization is to
use a pad and via design transparent to the
signal. This typically means using a small
diameter via with a surface-mount connector
and optimizing the clearance holes in
the planes. Alternatively, you could use a
copper fill adjacent to the signal via being
probed, with the copper fill connected to
return path vias adjacent to the signal via so
you could use microprobes.
Figure 1 shows the TDR response for
different connection designs. The top curve
is the TDR response (with a roughly 35 ps
rise time) for a conventional through-hole
Sub Miniature version A (SMA) connection to a bottom trace. On this scale, one
division is a reflection coefficient of 10%
and corresponds to an impedance change
of about 10 Ohms. At this rise time, the
impedance discontinuity is more than 18
Ohms, and is predominately capacitive.
You might think that avoiding the vias
will prevent the impedance discontinuity,
but just as many problems can be generated
by an edge-coupled SMA attached directly
to a surface trace. The second curve in
Figure 1 shows the measured TDR response
of an edge-coupled launch using an SMA.
The impedance discontinuity is more than
18 Ohms at this rise time and is inductive.
One way to avoid this problem is to use
microprobes and design the surface pads
for probing. The key feature is to use a copper
fill shorted to all adjacent ground vias.
In Figure 1, the gray vias have been shorted
to the copper fill. With this configuration,
you can probe every signal.
The third TDR curve in Figure 1 shows
the response of a microprobe launch into
an optimized 50 Ohm stripline. The
impedance discontinuity at this rise time is
less than 5 Ohms and is inductive.
Finally, it is possible to use an SMA connection
to a circuit board trace if it is optimized.
The bottom curve in Figure 1 shows
such a connection. Its impedance discontinuity,
less than 5 Ohms, compares to a
microprobe launch.
High-Bandwidth Measurements
All high-bandwidth measurements take
advantage of what is normally a problem
encountered by high-bandwidth signals:
reflections from impedance discontinuities.
As a signal propagates down an interconnect,
if the instantaneous impedance the signal
sees ever changes, a reflection will occur
and the transmitted signal will be distorted.
The magnitude of the reflected signal will
depend on the change in impedance.
By using a calibrated reference signal –
a sine wave in the frequency domain and a
Gaussian step edge in the time domain –
and measuring the amount of signal
reflected back from an interconnect as well
as transmitted through it, you can extract
the electrical properties of the interconnect.
All of the electrical properties of the
interconnect path are contained in these
two basic measurements.
When displaying data in the frequency
domain, the reflected signal is called the
return loss and the transmitted signal is
called the insertion loss. These two metrics
have become the universal standard to characterize
the fundamental properties of an
interconnect, such as a channel path in a
backplane. Many of the important physical
layer properties of a backplane can be read
directly from the return and insertion loss of
both single-ended and differential channels.
When displaying data in the time
domain, the reflected signal gives direct
insight into how the physical structure
contributes to electrical impedance discontinuities.
The transmitted signal in the
time domain gives a direct measure of the
propagation delay and rise time degradation.
From this result, an eye diagram can
be synthesized.
Whether you’ve measured the data in
the time or frequency domain, it can be
transformed into either one. A VNA will
measure the response in the frequency
domain, while a TDR will measure the
response in the time domain. With appropriate
software, you can convert the data
from either instrument into both domains.
All high-speed serial links today use
differential signaling and backplane channels
routed on differential pairs. For these
structures, the same metrics of return and
insertion loss are used, but there are additional
terms. Both differential and common
signals will have a return and
insertion loss, with mode conversion
terms of differential signal in, common
signal out and common signal in, and differential
signal out.
Differential S-Parameters
The description of return and insertion loss
measurements borrows from a formalism
heavily used in the RF world based on scattering
or S-parameters. It’s just a shorthand
way of keeping track of all the different
measurements.
In a differential channel, the interconnect
is a single, differential pair, with the two ends
labeled port 1 and port 2. The ratio of the
reflected sine wave signal coming out of port
1 to the incident sine wave signal going into
port 1 is labeled S11. This is the return loss.
The ratio of the transmitted sine wave
signal coming out of port 2 to the incident
sine wave signal going into port 1 is labeled
S21. This is the insertion loss.
A complication arises in a differential
pair, where you must consider not only the
port at which signals appear but also the
nature of the signal (differential or common).
There are four choices:
- A differential signal going in and coming
out, which would be the differential
return and insertion loss, SDD11
and SDD21
- A common signal going in and coming
out, which would be the common
return and insertion loss, SCC11 and
SCC21
- A differential signal going in and a
common signal coming out, a type of
mode conversion, SCD11 and SCD21
- A common signal going in and a differential
signal coming out, a type of
mode conversion, SDC11 and SDC21.
Don’t forget the case of the signal going
in from port 2 rather than port 1. All of
these combinations result in 16 differential
S-parameters, which are arrayed in a
matrix. Each set of terms has significance,
but the most important are the differential
return and insertion loss and the differential
to common mode conversion.
Differential Return Loss
SDD11 is a direct measure of the impedance
discontinuities encountered by the
differential signal propagating through
the channel. Figure 2 is an example of the
measured differential return loss of a
backplane trace in the frequency domain
up to 20 GHz. The more negative the
decibel value, the less reflected signal and
the better the impedance match.
It’s a little difficult to interpret the measurement
in the frequency domain. This is a
case where transforming the data to the
time domain gives immediate insight.
Figure 3 is the same data displayed in
the time domain. In this display, you can
identify the discontinuity from the SMA
launch, the high impedance of the daughtercard,
and the capacitive discontinuity of
the vias in the backplane.
Differential Insertion Loss
SDD21 is a direct measure of the quality
of the transmitted differential signal
through the channel. In the frequency
domain we can read the bandwidth of the
interconnect directly off the screen. The
maximum useable bandwidth of the channel
is set by the frequency at which the
attenuation is below the usable value, typically
about -15 dB of loss, depending on
the SerDes. The more discontinuities and
losses, the higher the attenuation, and the
lower the bandwidth.
Figure 4 shows the measured SDD21
for two different length channels, including
the higher bandwidth of the shorter
channel.
Using the limiting attenuation as -15
dB, the short channel has a usable bandwidth
of about 4 GHz, and the long channel
has a usable bandwidth of about 3 GHz.
This would correspond to a usable bit rate
of roughly 2.5 Gbps and 2 Gbps. However,
it is more than just the attenuation that
determines the maximum usable bit rate.
A better estimator for the maximum
usable bit rate is the eye diagram. Even
though this differential insertion loss was
measured in the frequency domain, it can
be translated into the time domain, and as
a response function can be used to calculate
an eye diagram.
Figure 5 shows the calculated eye diagram
for a 25-inch channel with 2.5 Gbps
and 5 Gbps signals. Based on this measured
response, this channel might be useful
for even 5 Gbps data rates, with an
appropriate receiver.
Mode Conversion
Any asymmetry between the two lines that
make up the differential pair will convert
some of the transmitted differential signal
into common signal. This will create two
problems. If any of this created common
signal gets out of the channel onto external
twisted pairs, it will potentially contribute
to electromagnetic interference. Of course,
every good design should have integrated
common signal chokes in all external twisted
pair connectors. However, it is always
good practice to try to reduce the source of
the noise before filtering.
The second problem isn’t so much from
the common signal created but from the
impact on the differential signal from what
caused the conversion. One of the most
common sources of mode conversion is a
difference in the time delay of each channel.
This line-to-line skew within a channel
will convert differential signals to common
signals and result in increased rise time
degradation of the differential signal and
larger deterministic jitter.
The total amount of common signal
coming out of port 2, based on a pure differential
signal going into port 1, is
described by the SCD21 term. Figure 6
shows the response for this channel.
Looking at the time evolution of the
creation of the converted common signal
coming out of port 1, we can gain insight
into where the conversion might be
occurring. Figure 7 shows the SCD11
term, displayed in the time domain, compared
with the SDD11 term, which has
information about the physical features of
the channel.
It appears as though most of the mode
conversion occurs in the via field of the
backplane side of the connector to the
daughtercard. Additional mode conversion
exists at each of the connector locations in
the backplane. This might be caused by the
via fields or an asymmetry between the two
lines in the differential pair, such as a spatial
difference in the dielectric constant
each trace sees.
Conclusion
Everything you ever wanted to know about
the electrical characteristics of a differential
channel is contained in the differential Sparameters.
They can be measured in the
time domain or the frequency domain and
displayed in either, and each one offers a
different insight.
Measurements play an important role in
risk reduction when designing systems
incorporating Rocket IO or RocketIO X
transceivers. Although it is important to
integrate simulation tools into the design
process to perform cost/benefit analyses of
technology and design tradeoffs, it is also
important to use measurements to verify
the accuracy of the simulation process.
Measurements can also offer immediate
insight into the behavior of first article
hardware to evaluate whether they meet
specifications, and how well the interconnects
will interact with the silicon.
Additional Resources
For more information about this
and other signal integrity topics, visit www.BogEnt.com.
Acknowledgments
The data in this paper was graciously
provided by Maria Brown of Agilent
Technologies and Al Neves and Dima
Smolyansky of TDA Systems Inc.
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